Chebyshev filter

Chebyshev filters are analog or digital filters that have a steeper roll-off than Butterworth filters, and have either passband ripple (type I) or stopband ripple (type II). Chebyshev filters have the property that they minimize the error between the idealized and the actual filter characteristic over the operating frequency range of the filter, but they achieve this with ripples in the passband. This type of filter is named after Pafnuty Chebyshev because its mathematical characteristics are derived from Chebyshev polynomials. Type I Chebyshev filters are usually referred to as "Chebyshev filters", while type II filters are usually called "inverse Chebyshev filters". Because of the passband ripple inherent in Chebyshev filters, filters with a smoother response in the passband but a more irregular response in the stopband are preferred for certain applications.

Type I Chebyshev filters (Chebyshev filters)

The frequency response of a fourth-order type I Chebyshev low-pass filter with $\varepsilon =1$

Type I Chebyshev filters are the most common types of Chebyshev filters. The gain (or amplitude) response, $G_{n}(\omega )$, as a function of angular frequency $\omega$ of the $n$th-order low-pass filter is equal to the absolute value of the transfer function $H_{n}(s)$ evaluated at $s=j\omega$:

$G_{n}(\omega )=\left|H_{n}(j\omega )\right|={\frac {1}{\sqrt {1+\varepsilon ^{2}T_{n}^{2}(\omega /\omega _{0})}}}$

where $\varepsilon$ is the ripple factor, $\omega _{0}$ is the cutoff frequency and $T_{n}$ is a Chebyshev polynomial of the $n$th order.

The passband exhibits equiripple behavior, with the ripple determined by the ripple factor $\varepsilon$. In the passband, the Chebyshev polynomial alternates between -1 and 1 so the filter gain alternate between maxima at $G=1$ and minima at $G=1/{\sqrt {1+\varepsilon ^{2}}}$.

The ripple factor ε is thus related to the passband ripple δ in decibels by:

$\varepsilon ={\sqrt {10^{\delta /10}-1}}.$

At the cutoff frequency $\omega _{0}$ the gain again has the value $1/{\sqrt {1+\varepsilon ^{2}}}$ but continues to drop into the stopband as the frequency increases. This behavior is shown in the diagram on the right. The common practice of defining the cutoff frequency at −3 dB is usually not applied to Chebyshev filters; instead the cutoff is taken as the point at which the gain falls to the value of the ripple for the final time.

The 3 dB frequency $\omega _{H}$ is related to $\omega _{0}$ by:

$\omega _{H}=\omega _{0}\cosh \left({\frac {1}{n}}\cosh ^{-1}{\frac {1}{\varepsilon }}\right).$

The order of a Chebyshev filter is equal to the number of reactive components (for example, inductors) needed to realize the filter using analog electronics.

An even steeper roll-off can be obtained if ripple is allowed in the stopband, by allowing zeros on the $\omega$-axis in the complex plane. While this produces near-infinite suppression at and near these zeros (limited by the quality factor of the components, parasitics, and related factors), overall suppression in the stopband is reduced. The result is called an elliptic filter, also known as a Cauer filter.

Poles and zeroes

Log of the absolute value of the gain of an 8th-order Chebyshev type I filter in complex frequency space (s = σ + jω) with ε = 0.1 and $\omega _{0}=1$. The white spots are poles and are arranged on an ellipse with a semi-axis of 0.3836... in σ and 1.071... in ω. The transfer function poles are those poles in the left half plane. Black corresponds to a gain of 0.05 or less, white corresponds to a gain of 20 or more.

For simplicity, it is assumed that the cutoff frequency is equal to unity. The poles $(\omega _{pm})$ of the gain function of the Chebyshev filter are the zeroes of the denominator of the gain function. Using the complex frequency $s$, these occur when:

$1+\varepsilon ^{2}T_{n}^{2}(-js)=0.\,$

Defining $-js=\cos(\theta )$ and using the trigonometric definition of the Chebyshev polynomials yields:

$1+\varepsilon ^{2}T_{n}^{2}(\cos(\theta ))=1+\varepsilon ^{2}\cos ^{2}(n\theta )=0.\,$

Solving for $\theta$

$\theta ={\frac {1}{n}}\arccos \left({\frac {\pm j}{\varepsilon }}\right)+{\frac {m\pi }{n}}$

where the multiple values of the arc cosine function are made explicit using the integer index $m$. The poles of the Chebyshev gain function are then:

$s_{pm}=j\cos(\theta )\,$

$=j\cos \left({\frac {1}{n}}\arccos \left({\frac {\pm j}{\varepsilon }}\right)+{\frac {m\pi }{n}}\right).$

Using the properties of the trigonometric and hyperbolic functions, this may be written in explicitly complex form:

$s_{pm}^{\pm }=\pm \sinh \left({\frac {1}{n}}\mathrm {arsinh} \left({\frac {1}{\varepsilon }}\right)\right)\sin(\theta _{m})$

$+j\cosh \left({\frac {1}{n}}\mathrm {arsinh} \left({\frac {1}{\varepsilon }}\right)\right)\cos(\theta _{m})$

where $m=1,2,...,n$ and

$\theta _{m}={\frac {\pi }{2}}\,{\frac {2m-1}{n}}.$

This may be viewed as an equation parametric in $\theta _{n}$ and it demonstrates that the poles lie on an ellipse in $s$-space centered at $s=0$ with a real semi-axis of length $\sinh(\mathrm {arsinh} (1/\varepsilon )/n)$ and an imaginary semi-axis of length of $\cosh(\mathrm {arsinh} (1/\varepsilon )/n).$

The transfer function

The above expression yields the poles of the gain $G$. For each complex pole, there is another which is the complex conjugate, and for each conjugate pair there are two more that are the negatives of the pair. The transfer function must be stable, so that its poles are those of the gain that have negative real parts and therefore lie in the left half plane of complex frequency space. The transfer function is then given by

$H(s)={\frac {1}{2^{n-1}\varepsilon }}\ \prod _{m=1}^{n}{\frac {1}{(s-s_{pm}^{-})}}$

where $s_{pm}^{-}$ are only those poles of the gain with a negative sign in front of the real term, obtained from the above equation.

The group delay

Gain and group delay of a 5th-order type I Chebyshev filter with ε = 0.5.

The group delay is defined as the derivative of the phase with respect to angular frequency:

$\tau _{g}=-{\frac {d}{d\omega }}\arg(H(j\omega ))$

The gain and the group delay for a 5th-order type I Chebyshev filter with ε=0.5 are plotted in the graph on the left. It's stop band has no ripples. But the ripples of group delay in its passband indicate that different frequency components have different delay, which along with the ripples of gain in its passband results in distortion of the waveform's shape.

Even order modifications

Even order Chebyshev filters implemented with passive elements, typically inductors, capacitors, and transmission lines, with terminations of equal value on each side cannot be implemented with the traditional Chebyshev transfer function. This is due to the physical inability to accommodate the even order Chebyshev reflection zeros that result in a scattering matrix S12 values that exceed the S12 value at $\omega =0$. If it is not feasible to design the filter with one of the terminations increased or decreased to accommodate the pass band S12, then the Chebyshev transfer function must be modified so as to move the lowest even order reflection zero to $\omega =0$ while maintaining the equi-ripple response of the pass band.

The needed modification involves mapping each pole of the Chebyshev transfer function in a manner that maps the lowest frequency reflection zero to zero and the remaining poles as needed to maintain the equi-ripple pass band. The lowest frequency reflection zero may be found from the Chebyshev Nodes, $cos{\Bigl (}{\frac {\pi (n-1)}{2n}}{\Bigl )}$. The complete Chebyshev pole mapping function is shown below.

$P'=\left[{\sqrt {\left({\frac {P^{2}+cos^{2}{\Bigl (}{\frac {\pi (n-1)}{2n}}{\Bigl )}}{1-{cos^{2}{\Bigl (}{\frac {\pi (n-1)}{2n}}{\Bigl )}}}}\right)}}\right]_{\text{Left Half Plane }}$

Where:

n is the order of the filter (must be even)

P is a traditional Chebyshev transfer function pole

P' is the mapped pole for the modified even order transfer function.

"Left Half Plane" indicates to use the square root containing a negative real value.

When complete, a replacement equi-ripple transfer function is created with reflection zero scattering matrix values for S12 of one and S11 of zero when implemented with equally terminated passive networks. The illustration below shows an 8th order Chebyshev filter modified to support even order equally terminated passive networks by relocating the lowest frequency reflection zero from a finite frequency to 0 while maintaining an equi-ripple pass band frequency response.

Even order modified Chebyshev illustration

The LC element value formulas in the Cauer topology are not applicable to the even order modified Chebyshev transfer function, and cannot be used. It is therefore necessary to calculate the LC values from traditional continued fractions of the impedance function, which may be derived from the reflection coefficient, which in turn may be derived from the transfer function.

Setting the cutoff attenuation

Pass band cutoff attenuation for Chebyshev filters is usually the same as the pass band ripple attenuation, set by the computation above. However, many applications such as diplexers and triplexers, require a cutoff attenuation of -3.0103 dB in order to obtain the needed reflections. Other specialized applications may require other specific values for cutoff attenuation for various reasons. It is therefore useful to have a means available of setting the Chebyshev pass band cutoff attenuation independently of the pass band ripple attenuation, such as -1 dB, -10 dB, etc. The cutoff attenuation may be set by frequency scaling the poles of the transfer function.

The scaling factor may be determined by direct algebraic manipulation of the defining Chebyshev filter function, $G_{n}(\omega )$, including $\varepsilon$ and $T_{n}(\omega /\omega _{0})$. The general definition of the Chebyshev function, $T_{n}(\omega /\omega _{0})=cos(n\cos ^{-1}(\omega /\omega _{0}))$ is required, which may be derived from the Chebyshev Polynomials equations, and the inverse Chebyshev function, $T_{n}^{-1}(\omega /\omega _{0})=cos(\cos ^{-1}(\omega /\omega _{0})/n)$. To keep the numbers real for values of $\omega /\omega _{0}\geq 1$, complex hyperbolic identities may be used to rewrite the equations as, $T_{n}(\omega /\omega _{0})=cosh(n\cosh ^{-1}(\omega /\omega _{0}))$ and $T_{n}^{-1}(\omega /\omega _{0})=cosh(\cosh ^{-1}(\omega /\omega _{0})/n)$.

Using simple algebra on the above equations and references, the expression to scale each Chebyshev poles is:

${\begin{aligned}p_{A}&=p_{1}/T_{n}^{-1}{\Biggr (}{\sqrt {\frac {10^{{-\alpha }/10}-1}{10^{-\delta /10}-1}}},n{\Biggr )}\qquad &{\text{For }}0>\delta >-\infty {\text{ and }}\delta \geq \alpha >-\infty \\&=p_{1}*sech{\Biggr (}{\frac {1}{n}}cosh^{-1}{\Bigr (}{\sqrt {\frac {10^{-\alpha /10}-1}{10^{-\delta /10}-1}}}{\Bigr )}{\Biggr )}&{\text{For }}0>\delta >-\infty {\text{ and }}\delta \geq \alpha >-\infty \\\end{aligned}}$

Where:

$p_{A}$ is the relocated pole positioned to set the desired cutoff attenuation.

$p_{1}$ is a ripple cutoff pole that lies on the oval.

$\delta$ is the passband attenuation ripple in dB (-.05 dB, -1 dB, etc.)).

$\alpha$ is the desired passband attenuation at the cutoff frequency in dB (−1 dB, -3 dB, −10 dB, etc.)

$n$ is the number of poles (the order of the filter).

A quick sanity check on the above equation using passband ripple attenuation for the passband cutoff attenuation $(\alpha =\delta )$ reveals that the pole adjustment will be 1.0 for this case, which is what is expected.

Even order modified cutoff attenuation adjustment

For Chebyshev filters being designed with modified for even order pass band ripple for passive equally terminated filters, the attenuation frequency computation needs to include the even order adjustment by performing the even order adjustment operation on the computed attenuation frequency. This makes the even order adjustment arithmetic slightly simpler, since frequency can be treated as a real variable, in this case $((J\omega )^{2}{\text{ becomes }}-\omega ^{2})$.

${\begin{aligned}p_{A}=p_{1}{\sqrt {\frac {1-{cos^{2}({\frac {\pi (n-1)}{2n}})}}{cosh^{2}{\Biggr (}{\frac {1}{n}}cosh^{-1}{\Bigr (}{\sqrt {\frac {10^{-\alpha /10}-1}{10^{-\delta /10}-1}}}{\Bigr )}{\Biggr )}-cos^{2}({\frac {\pi (n-1)}{2n}})}}}{\text{ For }}0>\delta >-\infty {\text{ and }}\delta \geq \alpha >-\infty \\\end{aligned}}$

Where:

$p_{A}$ is the relocated pole positioned to set the desired cutoff attenuation.

$p_{1}$ is a ripple cutoff pole that has been modified for even order pass bands.

$\delta$ is the passband attenuation ripple in dB (-.05 dB, -1 dB, etc.)).

$\alpha$ is the desired passband attenuation at the cutoff frequency in dB (−1 dB, -3 dB, −10 dB, etc.)

$n$ is the number of poles (the order of the filter).

$cos({\frac {\pi (n-1)}{2n}})$ is the smallest even order Chebyshev Node

Chebyshev transmission zeros

Chebyshev filters may be designed with arbitrarily placed finite transmission zeros in the stop band while retaining an equi-ripple pass band. Stop band zeros along the $j\omega$ axis are generally used to eliminate unwanted frequencies. Stop band zeros along the real axis or quadruplet stop band zeros in the complex plane may be used to modify the group delay to a more desirable shape. The transmission zeros design utilizes characteristic polynomials, K(S), to place the transmission and reflection zeros, which in turn are used to create the transfer function, $G(s)$,

$G(s)={\sqrt {\frac {1}{1+\varepsilon ^{2}K(s)K(-s)}}}{\bigg |}_{\text{left half plane (LHP) poles}}$

The calculation of K(S) relies upon the following observed equality.

${\begin{aligned}&{\begin{array}{lcr}&{\bigg |}\prod _{i=1}^{N}{\frac {j\omega {\frac {\sqrt {z_{i}^{2}+1}}{z_{i}}}+{\sqrt {1-\omega ^{2}}}}{(1-j\omega /z_{i})}}{\bigg |}=1.&&{\text{ for }}0\leq \omega \leq 1\end{array}}\\\end{aligned}}$

for all $z_{i}=\infty$, imaginary conjugate pairs, quadruplet conjugate pairs, or real opposing signed pairs.

Given the magnitude is always one in the pass bane ($0\leq \omega \leq 1$) the rational and irrational terms must vary between 0 and 1. Therefore, if only the rational term is used to create the $K(s)$ characteristic function, an equi-ripple response is expected in the pass band, and characteristic poles (transmission zeros) are expected at all $s=z_{i}$.

The design process for K(S) using the above expression is below.

${\begin{aligned}K(s)&={\frac {{\bigg \{}\prod _{i=1}^{N}{\bigg (}M_{i}s+{\sqrt {s^{2}+1}}{\bigg )}{\bigg \}}_{\text{rational term only}}}{\prod _{i=1}^{N}(1-s/z_{i})}}\\M_{i}&={\frac {\sqrt {z_{i}^{2}+1}}{z_{i}}}{\text{ for }}\sigma _{i}\neq 0{\text{ or }}\omega _{i}>1\\&={\frac {\sqrt {\omega _{i}^{2}-1}}{\omega _{i}}}{\text{ for }}\sigma _{i}=0{\text{ and }}\omega _{i}>1\\&=1{\text{ for }}\omega _{i}=\infty \\z_{i}&=\sigma _{i}+j\omega _{i}={\text{ complex transmision zero}}\\\end{aligned}}$

Use the positive $M_{i}$ solution for real and imaginary $z_{i}$ pairs. Use the positive real and conjugate imaginary $M_{i}$solution for quadruplet complex $z_{i}$ pairs.

$K(s)$ should be normalized such that $|K(s)|=1{\text{ at }}s=j$, if needed.

The, "rational terms only" indicates to keep the rational part of the product, and to discard the irrational part. The rational term may be obtained by manually performing the polynomial arithmetic, or with the short cut below which is a solution derived from polynomial arithmetic and uses binomial coefficients..

${\begin{aligned}&B=\prod _{i=1}^{N}(M_{i}s+1)\\&K(s)_{num}=\sum _{i=N}^{i\geq 0{\text{, step }}=-2}{\bigg [}\sum _{j=i}^{j\geq 0{\text{, step }}=-2}B_{j}{\binom {(N-j)/2}{(N-i)/2}}{\bigg ]}s^{i}\\&N={\text{ order of the Chebyshev filter}}\\&B={\text{a polynomial created by the product of the specified factors}}\\&B_{j}={\text{ the }}j_{th}{\text{ order coefficent of polynomial }}B\\&{\binom {n}{k}}{\text{ is the binomial coefficient function}}\\\end{aligned}}$

When all M values are set to one, then $K(s)_{num}$ will be the standard Chebyshev equation, which is expected since the all transmission zeros are it $\infty$. It is also important to remember that even order finite transmission zero Chebyshev filters have the same limitation as the all-pole case in that they cannot be constructed using equally terminated passive networks. The same even order modification may be made to the even order characteristic polynomials, $K(s)$, to make equally terminated passive network implementations possible. However, the even order modification will also move the finite transmission zeros slightly. This movement may be significantly mitigated by propositioning the transmission zeros with the inverse of the even order modification using the lowest Chebyshev node, $cos(\pi (N-1)/(2N))$.

${\begin{aligned}&z_{i}'={\sqrt {z_{i}^{2}(1.-C_{0}^{2})-C_{0}^{2}}}\\&C_{0}^{2}=cos^{2}({\frac {\pi (N-1)}{2N}})\\&z_{i}={\text{desired finite transmisison zero}}\\&z_{i}'={\text{prepositioned finite transmisison zero}}\\\end{aligned}}$

Simple example

Design a 3 pole Chebyshev filter with a 1 dB pass band, a transmission zero at 2 rad/sec, and a transmission zero at $\infty$:

${\begin{aligned}&M_{1}=M_{2}={\sqrt {(j2)^{2}+1}}/j2={\sqrt {1-4}}/j2={\sqrt {3}}/2=0.866025{\text{, }}M_{3}={\sqrt {\infty ^{2}+1}}/\infty =1\\&\\&{\text{Full polynomial derivation:}}\\&K(s)_{num}={\bigr (}0.86602540s+{\sqrt {s^{2}+1}}{\bigr )}{\bigr (}0.86602540s+{\sqrt {s^{2}+1}}{\bigr )}{\bigr (}s+{\sqrt {s^{2}+1}}{\bigr )}\\&K(s)_{num}=3.4820508s^{3}+2.7320508s+{\sqrt {\dots }}\\&{\text{discarding the irrational }}{\sqrt {\dots }}{\text{ and keeping only the rational part:}}\\&K(s)_{num}=3.4820508s^{3}+2.7320508s\\&\\&K(s)_{num}{\text{ shortcut derivation:}}\\&B=(0.86602540s+1)(0.86602540s+1)(s+1)=.75s^{3}+2.4820508s^{2}+2.7320508s+1\\&K(s)_{num}={\bigg (}0.75{\binom {0}{0}}+2.4820508{\binom {1}{0}}{\bigg )}s^{3}+{\bigg (}2.7320508{\binom {1}{1}}{\bigg )}s\\&K(s)_{num}=3.4820508s^{3}+2.7320508s\\&\\&k(s)_{den}=({\frac {s}{j2}}+1)({\frac {s}{-j2}}+1)=0.25s^{2}+1\\&{\text{Check }}|K(s)|{\text{ at }}s=j{\text{ to insure it is unity, and adjust with a constant, if necessary:}}\\&{\bigg |}{\frac {K(s)_{num}(s=j)}{K(s)_{den}(s=j)}}{\bigg |}=1{\text{ Check!}}\\&K(s)={\frac {3.4820508s^{3}+2.7320508s}{0.25s^{2}+1}}\\\end{aligned}}$

To find the $G(s)$ transfer function, do the following.

${\begin{aligned}&\varepsilon ^{2}=10^{1dB/10.}-1.=.25892541\\&G(s)={\sqrt {G(s)G(-s)}}{\bigg |}_{\text{LHP poles}}={\sqrt {\frac {1}{1+\varepsilon ^{2}K(s)K(-s)}}}{\bigg |}_{\text{LHP poles}}={\sqrt {\frac {K(s)_{den}K(-s)_{den}}{1+\varepsilon ^{2}K(s)_{num}K(-s)_{num}}}}{\bigg |}_{\text{LHP poles}}\\&={\sqrt {\frac {\{0.25(s)^{2}+1\}\{{0.25(-s)^{2}+1}\}}{1+.25892541\{3.4820508(s)^{3}+2.7320508(s)\}\{3.4820508(-s)^{3}+2.7320508(-s)\}}}}{\bigg |}_{\text{LHP poles}}\\&={\frac {0.25(s)^{2}+1}{{\sqrt {-3.1393872s^{6}-4.8638872s^{4}-1.4326456s^{2}+1}}{\bigr |}_{\text{LHP poles}}}}\\\end{aligned}}$

To obtain $G(s)$ from the left half plane, factor the numerator and denominator to obtain the roots. Discard all roots from the right half plane of the denominator, half the repeated roots in the numerator, and rebuild $G(s)$ with the remaining roots. Generally, normalize $|G(s)|$ to 1 at $s=0$.

${\begin{aligned}&G(s)={\frac {0.25s^{2}+1}{1.7718316s^{3}+1.7200107s^{2}+2.2074118s+1}}\\\end{aligned}}$

To confirm that the example $G(s)$ is correct, the plot of $G(s)$ along $j\omega$ is shown below with a pass band ripple of 1 dB, a cut off frequency of 1 rad/sec, and a stop band zero at 2 rad/sec.

Chebyshev transmission zero at 2 rad/sec

Asymmetric band pass filter

Chebyshev band pass filters may be designed with a geometrically asymmetric frequency response by placing the desired number of transmission zeros at zero and infinity with the use of the more generalized form of the Chebyshev transmission zeros equation above, and shown below. The $K(s)$ equations below consider a frequency normalized pass band from 1 to $\omega _{2}$. If the number of transmission zeros at 0 is not the same as the number of transmission zeros at $\infty$, the filter will be geometrically asymmetric. The filter will also be asymmetric if finite transmission zeros are not place symmetrically about the geometric center frequency, which in this case is ${\sqrt {\omega _{2}}}$. There is a restriction in that he filter must be net even order, that is the sum of all the poles must be even, to make the asymmetric $K(s)$ equation produce usable results. Real and complex quadruplet transmission zeros may also be created using this technique and are useful to modify the group delay response, just as in the low pass case. The derivation of the characteristic equation, $K(s)$, to create an asymmetric Chebyshev band pass filter is shown below.

${\begin{aligned}K(s)&={\frac {{\bigg \{}\prod _{i=1}^{N}{\bigg (}M_{i}{\sqrt {s^{2}+\omega _{2}^{2}}}+{\sqrt {s^{2}+1}}{\bigg )}{\bigg \}}_{\text{rational term only}}}{s^{N_{z}}\prod _{i=1}^{N_{f}}(1-s/z_{i})}}\\M_{i}&={\sqrt {\frac {z_{i}^{2}+1}{z_{i}^{2}+\omega _{2}^{2}}}}{\text{ for }}\sigma _{i}\neq 0{\text{ or }}\omega _{i}<1{\text{ or }}\omega _{i}>\omega _{2}\\&={\sqrt {\frac {1-\omega _{i}^{2}}{\omega _{2}^{2}-\omega _{i}^{2}}}}{\text{ for }}\sigma _{i}=0{\text{ and }}0<\omega _{i}<1\\&={\sqrt {\frac {\omega _{i}^{2}-1}{\omega _{i}^{2}-\omega _{2}^{2}}}}{\text{ for }}\sigma _{i}=0{\text{ and }}\omega _{2}<\omega _{i}<\infty \\&={\frac {1}{\omega _{2}}}{\text{ for }}z_{i}=0\\&=1{\text{ for }}z_{i}=\infty \\N_{z}&={\text{ number of transmission zeros at zero}}\\N_{f}&={\text{ number of finite transmission zeros (imaginay, real, and complex)}}\\z_{i}&=\sigma _{i}+j\omega _{i}={\text{ complex transmision zero}}\\\omega _{2}&={\text{ upper passband corner frequency (lower corner is normalized to 1)}}\\\end{aligned}}$

$K(s)$ should be normalized such that $|K(s)|=1{\text{ at }}s=j$, if needed.

Simple example

Design an asymmetric Chebyshev filter with 1dB pass band ripple from 1 to 2 rad/sec, one transmission zero at $\infty$, and three transmission zeros at 0. By applying the numeral values to the equations above, the characteristic polynomials, $K(s)$, may be calculated as follows.

${\begin{aligned}\omega _{2}&=2\\M_{1}&=M_{2}=M_{3}=.5\\M_{4}&=1\\K(s)&={\frac {{\bigg \{}{\bigg (}.5{\sqrt {s^{2}+2^{2}}}+{\sqrt {s^{2}+1}}{\bigg )}{\bigg (}.5{\sqrt {s^{2}+2^{2}}}+{\sqrt {s^{2}+1}}{\bigg )}{\bigg (}.5{\sqrt {s^{2}+2^{2}}}+{\sqrt {s^{2}+1}}{\bigg )}{\bigg (}{\sqrt {s^{2}+2^{2}}}+{\sqrt {s^{2}+1}}{\bigg )}{\bigg \}}_{\text{rational term only}}}{s^{3}}}\\K(s)&=C{\frac {3.375s^{4}+14.25s^{2}+12+{\sqrt {\dots }}}{s^{3}}}{\text{ where C is a constant used to normalize the magnitude to 1 at }}s=j\\\end{aligned}}$

Discarding the irrational part and normalizing $|K(s)|$ to 1 at s=j:

${\begin{aligned}K(s)&={\frac {s^{4}+12.666667s^{2}+10.666667}{s^{3}}}\\\end{aligned}}$

Use the same process as in the low pass case to find $G(s)$ from $K(s)$, using constant $C$ to scale the magnitude.

${\begin{aligned}\varepsilon ^{2}&=10^{1dB/10.}-1.=.25892541\\G(s)&=C{\sqrt {\frac {K(s)_{den}K(-s)_{den}}{K(s)_{den}K(-s)_{den}+\varepsilon ^{2}K(s)_{num}K(-s)_{num}}}}{\bigg |}_{\text{LHP poles}}\\&=C{\frac {s^{3}}{{\sqrt {(s)^{3}(-s)^{3}+.25892541\{(s)^{4}+12.666667(s)^{2}+10.666667\}\{(-s)^{4}+12.666667(-s)^{2}+10.666667\}}}{\bigr |}_{\text{LHP poles}}}}\\&=C{\frac {s^{3}}{{\sqrt {(s)^{3}(-s)^{3}+.25892541\{(s)^{4}+12.666667(s)^{2}+10.666667\}\{(-s)^{4}+12.666667(-s)^{2}+10.666667\}}}{\bigr |}_{\text{LHP poles}}}}\\\end{aligned}}$

When reconstructing the denominator from the left half plane poles, it will be necessary to set the $G(s)$ magnitude such that the reflection zeros occur at 0dB. To do this, $G(s)$ should be scaled such that $|G(s)|$ = -1dB at the pass band corner frequencies, $s=j$ and $s=j2$. Once accomplished, the final transfer function for the designed asymmetric Chebyshev filter is shown below.

${\begin{aligned}G(s)&={\frac {0.18424001s^{3}}{0.28125000s^{4}+0.34089984s^{3}+1.3337548s^{2}+0.54084155s+1}}\\\end{aligned}}$

Evaluating $|G(s)|$ at s=j and at s=2j produces a value of -1dB in both cases, yielding an assurance that the example has been synthesized correctly. The frequency response is below, showing a Chebyshev 1dB equi-ripple pass band response for $1<\omega <2$, cutoff attenuation of -1dB at the pass band edges, -60dB / decade attenuation toward $\omega =0$, -20dB / decade attenuation toward $\omega =\infty$, and Chebyshev style steepened slopes near the pass band edges.

Simulation showing asymmetric Chebyshev

Type II Chebyshev filters (inverse Chebyshev filters)

The frequency response of a fifth-order type II Chebyshev low-pass filter with $\varepsilon =0.01$

Also known as inverse Chebyshev filters, the Type II Chebyshev filter type is less common because it does not roll off as fast as Type I, and requires more components. It has no ripple in the passband, but does have equiripple in the stopband. The gain is:

$G_{n}(\omega )={\frac {1}{\sqrt {1+{\frac {1}{\varepsilon ^{2}T_{n}^{2}(\omega _{0}/\omega )}}}}}={\sqrt {\frac {\varepsilon ^{2}T_{n}^{2}(\omega _{0}/\omega )}{1+\varepsilon ^{2}T_{n}^{2}(\omega _{0}/\omega )}}}.$

In the stopband, the Chebyshev polynomial oscillates between -1 and 1 so that the gain will oscillate between zero and

${\frac {1}{\sqrt {1+{\frac {1}{\varepsilon ^{2}}}}}}$

and the smallest frequency at which this maximum is attained is the cutoff frequency $\omega _{o}$. The parameter ε is thus related to the stopband attenuation γ in decibels by:

$\varepsilon ={\frac {1}{\sqrt {10^{\gamma /10}-1}}}.$

For a stopband attenuation of 5 dB, ε = 0.6801; for an attenuation of 10 dB, ε = 0.3333. The frequency f₀ = ω₀/2π is the cutoff frequency. The 3 dB frequency f_H is related to f₀ by:

$f_{H}={\frac {f_{0}}{\cosh \left({\frac {1}{n}}\cosh ^{-1}{\frac {1}{\varepsilon }}\right)}}.$

Poles and zeroes

Log of the absolute value of the gain of an 8th order Chebyshev type II filter in complex frequency space (s=σ+jω) with ε = 0.1 and $\omega _{0}=1$. The white spots are poles and the black spots are zeroes. All 16 poles are shown. Each zero has multiplicity of two, and 12 zeroes are shown and four are located outside the picture, two on the positive ω axis, and two on the negative. The poles of the transfer function are poles on the left half plane and the zeroes of the transfer function are the zeroes, but with multiplicity 1. Black corresponds to a gain of 0.05 or less, white corresponds to a gain of 20 or more.

Assuming that the cutoff frequency is equal to unity, the poles $(\omega _{pm})$ of the gain of the Chebyshev filter are the zeroes of the denominator of the gain:

$1+\varepsilon ^{2}T_{n}^{2}(-1/js_{pm})=0.$

The poles of gain of the type II Chebyshev filter are the inverse of the poles of the type I filter:

${\frac {1}{s_{pm}^{\pm }}}=\pm \sinh \left({\frac {1}{n}}\mathrm {arsinh} \left({\frac {1}{\varepsilon }}\right)\right)\sin(\theta _{m})$

$\qquad +j\cosh \left({\frac {1}{n}}\mathrm {arsinh} \left({\frac {1}{\varepsilon }}\right)\right)\cos(\theta _{m})$

where $m=1,2,...n$. The zeroes $(\omega _{zm})$ of the type II Chebyshev filter are the zeroes of the numerator of the gain:

$\varepsilon ^{2}T_{n}^{2}(-1/js_{zm})=0.\,$

The zeroes of the type II Chebyshev filter are therefore the inverse of the zeroes of the Chebyshev polynomial.

$1/s_{zm}=-j\cos \left({\frac {\pi }{2}}\,{\frac {2m-1}{n}}\right)$

for $m=1,2,...n$.

The transfer function

The transfer function is given by the poles in the left half plane of the gain function, and has the same zeroes but these zeroes are single rather than double zeroes.

The group delay

Gain and group delay of a fifth-order type II Chebyshev filter with ε = 0.1.

The gain and the group delay for a fifth-order type II Chebyshev filter with ε=0.1 are plotted in the graph on the left. It can be seen that there are ripples in the gain in the stopband but not in the pass band.

Even order modifications

Just like Chebyshev filter even order filters, the standard Chebyshev II even order filter cannot be implemented with equally terminated passive elements. In the Chebyshev Ii case, this is due to finite attenuation of S12 in the stop band. However, even order Chebyshev II filters may be modified by translating the highest frequency finite transmission zero to infinity, while maintaining the equi-ripple functions of the Chebyshev II stop band. To do this translation, an even order modified Chebyshev function is used in place of the standard Chebyshev function to define the Chebyshev II poles needed to create the even order modified Chebyshev II transfer function. Zeros are created using the roots of the even order modified Chebyshev polynomial, which are the even order modified Chebyshev nodes.

The illustration below shows an 8th order Inverse Chebyshev filter modified to support even order equally terminated passive networks by relocating the highest frequency transmission zero from a finite frequency to $\infty$ while maintaining an equi-ripple stop band frequency response.

Even order modified Inverse Chebyshev illustration

Setting the cutoff attenuation

The standard cutoff attenuation as described is the same at the pass band ripple attenuation. However, just as in Chebyshev filters, it is useful to set the cutoff attenuation to a desired value, and for the same reasons. Setting the Chebyshev II cutoff attenuation is the same as for Chebyshev cutoff attenuation, except the arithmetic attenuation and ripple entries are inverted in the equation and the poles and zeros are multiplied by the result, as opposed to divided by in the Chebyshev case..

${\begin{aligned}p_{A}&=p_{1}*T_{n}^{-1}{\Biggr (}{\sqrt {\frac {10^{{-\delta }/10}-1}{10^{-\alpha /10}-1}}},n{\Biggr )}\qquad &{\text{For }}0>\delta >-\infty {\text{ and }}0>\alpha \geq \delta \\&=p_{1}*cosh{\Biggr (}{\frac {1}{n}}cosh^{-1}{\Bigr (}{\sqrt {\frac {10^{-\delta /10}-1}{10^{-\alpha /10}-1}}}{\Bigr )}{\Biggr )}&{\text{For }}0>\delta >-\infty {\text{ and }}0>\alpha \geq \delta \\\end{aligned}}$

Even order modified cutoff attenuation adjustment

The same even order adjustment to the poles and zeros that was used for the Chebyshev even order modified cutoff attenuation may also be used for the Chebyshev II case, except the poles are multiplied by the result.

${\begin{aligned}p_{A}=p_{1}{\sqrt {\frac {cosh^{2}{\Biggr (}{\frac {1}{n}}cosh^{-1}{\Bigr (}{\sqrt {\frac {10^{-\delta /10}-1}{10^{-\alpha /10}-1}}}{\Bigr )}{\Biggr )}-cos^{2}({\frac {\pi (n-1)}{2n}})}{1-{cos^{2}({\frac {\pi (n-1)}{2n}})}}}}{\text{ For }}0>\delta >-\infty {\text{ and }}\delta \geq \alpha >-\infty \\\end{aligned}}$

Implementation

Cauer topology

A passive LC Chebyshev low-pass filter may be realized using a Cauer topology. The inductor or capacitor values of an $n$th-order Chebyshev prototype filter may be calculated from the following equations:

$G_{0}=1$

$G_{1}={\frac {2A_{1}}{\gamma }}$

$G_{k}={\frac {4A_{k-1}A_{k}}{B_{k-1}G_{k-1}}},\qquad k=2,3,4,\dots ,n$

$G_{n+1}={\begin{cases}1&{\text{if }}n{\text{ odd}}\\\coth ^{2}\left({\frac {\beta }{4}}\right)&{\text{if }}n{\text{ even}}\end{cases}}$

G₁, G_k are the capacitor or inductor element values. f_H, the 3 dB frequency is calculated with: $f_{H}=f_{0}\cosh \left({\frac {1}{n}}\cosh ^{-1}{\frac {1}{\varepsilon }}\right)$

The coefficients A, γ, β, A_k, and B_k may be calculated from the following equations:

$\gamma =\sinh \left({\frac {\beta }{2n}}\right)$

$\beta =\ln \left[\coth \left({\frac {\delta }{17.37}}\right)\right]$

$A_{k}=\sin {\frac {(2k-1)\pi }{2n}},\qquad k=1,2,3,\dots ,n$

$B_{k}=\gamma ^{2}+\sin ^{2}\left({\frac {k\pi }{n}}\right),\qquad k=1,2,3,\dots ,n$

where $\delta$ is the passband ripple in decibels. The number $17.37$ is rounded from the exact value $40/\ln(10)$.

Low-pass filter using Cauer topology

The calculated G_k values may then be converted into shunt capacitors and series inductors as shown on the right, or they may be converted into series capacitors and shunt inductors. For example,

• C_{1 shunt} = G₁, L_{2 series} = G₂, ...

or

• L_{1 shunt} = G₁, C_{1 series} = G₂, ...

Note that when G₁ is a shunt capacitor or series inductor, G₀ corresponds to the input resistance or conductance, respectively. The same relationship holds for G_n+1 and G_n. The resulting circuit is a normalized low-pass filter. Using frequency transformations and impedance scaling, the normalized low-pass filter may be transformed into high-pass, band-pass, and band-stop filters of any desired cutoff frequency or bandwidth.

Digital

As with most analog filters, the Chebyshev may be converted to a digital (discrete-time) recursive form via the bilinear transform. However, as digital filters have a finite bandwidth, the response shape of the transformed Chebyshev is warped. Alternatively, the Matched Z-transform method may be used, which does not warp the response.

Comparison with other linear filters

The following illustration shows the Chebyshev filters next to other common filter types obtained with the same number of coefficients (fifth order):

Chebyshev filters are sharper than the Butterworth filter; they are not as sharp as the elliptic one, but they show fewer ripples over the bandwidth.

See also

• Bessel filter
• Butterworth filter
• Chebyshev nodes
• Chebyshev polynomial
• Comb filter
• Elliptic filter
• Filter design

External links

• Media related to Chebyshev filters at Wikimedia Commons